Sponsored Links

 
Web LinkGrinder.com

Title: Variable inductor, oscillator including the variable inductor and radio terminal comprising this oscillator, and amplifier including the variable inductor and radio terminal comprising this am

Abstract: An amplifier comprises an amplifier circuit which comprises a first inductor as an impedance element for degeneration, and a control circuit which has a second inductor electro-magnetically connected to the first inductor, and changes a control current flowing through the second inductor to change an inductance value of the first inductor, thereby changing amplification characteristics of the amplifier circuit.

Patent Number: 7,417,501 Issued on 08/26/2008 to Fujimoto,   et al.

Inventors: Fujimoto; Ryuichi (Tokyo, JP), Itakura; Tetsuro (Tokyo, JP)
Assignee: Kabushiki Kaisha Toshiba (Tokyo, JP)
Appl. No.: 11/393,676
Filed: March 31, 2006
Related U.S. Patent Documents
Application NumberFiling DatePatent NumberIssue Date
11204166Aug., 20057190239
10448367May., 20037098737
Foreign Application Priority Data
May 31, 2002 [JP] 2002-160621
Jun 28, 2002 [JP] 2002-188946
Sep 18, 2002 [JP] 2002-270984

Current U.S. Class: 330/124R ; 330/295
Current International Class: H03F 3/68 (20060101)
Field of Search: 330/124R,295,84,126

References Cited [Referenced By]
U.S. Patent Documents
2334704 November 1943 Hilferty
4580114 April 1986 Upadhyayula
4833400 May 1989 Boutigny
5430411 July 1995 Boulic
5509078 April 1996 Hiraoka et al.
5912596 June 1999 Ghoshal
5994985 November 1999 Pehlke et al.
6121850 September 2000 Ghoshal
6392490 May 2002 Gramegna et al.
6642787 November 2003 Souetinov et al.
6750727 June 2004 Sutardja
Foreign Patent Documents
7-320942 Aug., 1995 JP
8-162331 Jun., 1996 JP
9-74319 Mar., 1997 JP
2000-223317 Aug., 2000 JP
2000-315919 Nov., 2000 JP
2002-9544 Jan., 2002 JP

Other References

Zhang, G. et al., "New Broadband Tunable Monolithic Microwave Floating Active Inductor." Electronic Letters, vol. 28, No. 1, pp. 78-81, Jan. 2, 1992. cited by other .
Fong, "Dual-Bank High-Linearity Variable-Gain Low-Noise Amplifiers for Wireless Applications," IEEE International Solid-State Circuits Conference (1999), pp. 224-225. cited by other .
Notification of Reasons for Rejection issued by the Japanese Patent Office, mailed Nov. 21, 2006, in Japanese Application No. 2003-150977 and English translation of Notification. cited by other.

Primary Examiner: Choe; Henry K
Attorney, Agent or Firm: Finnegan, Henderson, Farabow, Garrett & Dunner, L.L.P.
Parent Case Text


CROSS-REFERENCE TO RELATED APPLICATIONS

This is a division of Application No. 11/204,166, filed Aug. 16, 2005 (now U.S. Pat. No. 7,190,239), which is a division of application Ser. No. 10/448,367, filed May 30, 2003 (now U.S. Pat. No. 7,098,737), all of which are incorporated herein by reference. This application is also based upon and claims the benefit of priority from the prior Japanese Patent Applications No. 2002-160621, filed May 31, 2002; No. 2002-188946, filed Jun. 28, 2002; and No. 2002-270984, filed Sep. 18, 2002, the entire contents of all of which are incorporated herein by reference.
Claims


What is claimed is:

1. An amplifier comprising: an input terminal which receives an input signal; a variable gain amplification circuit configured to amplify the input signal with a variable gain of an amplification, and including a plurality of amplification stages connected to the input terminal in parallel to receive the input signal; and an input impedance adjustment circuit including a variable resistor circuit connected to the input terminal, configured to adjust a resistance value of the variable resistor circuit to compensate a change in input impedance corresponding to a change of the variable gain.

2. The amplifier according to claim 1, wherein the variable resistor circuit includes a plurality of variable resistors.

3. The amplifier according to claim 1, wherein at least one of the amplification stages has different amplification characteristics, one or more of the amplification stages being selectively operated to vary the gain of the gain amplification circuit.

4. The amplifier according to claim 1, wherein ones of the amplification stages have same amplification characteristics, a selected one or more of the amplification stages being selectively operated to vary the gain of the gain amplification circuit.

5. The amplifier according to claim 1, wherein the variable resistor circuit includes variable resistors connected in parallel and switches which are connected to the variable resistors, respectively.

6. The amplifier according to claim 1, wherein the plurality of amplification stages are directly connected to the input terminal, and the variable resistor circuit includes variable resistors connected in parallel and switches which are connected to the variable resistors, respectively.
Description


BACKGROUND OF THE INVENTION

1. Field of the Invention

This invention relates to a variable inductor, and more particularly to a variable inductor which uses a circuit constituted of an active element and a plurality of interconnected inductors to change inductance.

Additionally, this invention relates to an oscillator having an inductor, and a radio terminal having an inductor, and also relates to improvements in a circuit designing technology of an amplifier and a radio terminal and a gain varying method of the amplifier.

2. Description of the Related Art

Generally, in order to vary characteristics of an electronic circuit, characteristics of an active element or a value of a passive element included in the circuit is changed. For the active element, the characteristics of the active element can be changed by changing a bias voltage applied to the active element. For the passive element, a passive element, for example, a variable resistive element can be easily realized by using ON resistance of a MOSFET, and a variable capacitive element can be easily realized by using a pn junction.

For an inductor as a passive element, it is generally considered difficult to vary inductance while maintaining good characteristics. A method which uses an active element to constitute an inductor and varies inductance is disclosed in ELECTRONICS LETTERS 2nd January, Vol. 28, No. 21, pp. 78 to 80, 1992. However, there is a problem of bad characteristics of noise or distortion because of use of the active element for the inductor.

Thus, various technologies have been presented, which vary inductance without constituting an inductor of an active element. However, all of these technologies have problems in practical application. For example, Jpn. Pat. Appln. KOKAI Publication No. 8-162331 discloses a method which inserts a switch into the middle of an inductor, and turns the switch ON/OFF to change inductance. This method has a problem that ON resistance of the switch deteriorates performance of the variable inductor.

Jpn. Pat. Appln. KOKAI Publication No. 2000-223317 discloses a method which physically changes a shape of an inductor by a laser beam. This method necessitates a physical adjustment of the inductor after it is manufactured. Thus, there are problems of high manufacturing costs and a difficulty of changing inductance in a situation where a circuit is operated.

Jpn. Pat. Appln. KOKAI Publication No. 7-320942 discloses a technology which constitutes a variable inductor by using interconnection of a plurality of inductors. According to this method, a shape of the inductor is physically changed in order to change a coupling coefficient. Thus, there are problems of miniaturization and low costs of a circuit which constitutes the variable inductor.

Further, D. R. Pehlke et al. have presented, in U.S. Pat. No. 5,994,985, a technology which uses a directional coupler to separate an input signal into two, and controls amplitudes and phases of signals flowing through two interconnected inductors to change inductance. However, since the directional coupler is not generally suited for integration, there is a problem that it is difficult to realize a variable inductor by an integrated circuit.

By realizing the variable inductor, it is also possible to control even a voltage-controlled oscillator (VOC) which includes an LC resonant circuit (inductor: inductance L, capacitor: capacity C). An oscillation frequency (f) of the voltage controlled oscillator which includes this LC resonant circuit is generally represented by f=1/2[2.pi.(LC)1/2]. If the inductance l or the capacity C is controlled, the oscillation frequency (f) is controlled. However, in the conventional LC resonant circuit, it is difficult to realize a variable inductor as described above. Generally, therefore, the capacity C is varied, for example, a reverse bias application voltage to a pn junction diode is changed to vary the capacity C and, by changing this capacity, the oscillation frequency is changed.

If such a voltage controlled oscillator is formed as an integrated circuit on a semiconductor substrate, i.e., IC formation, a parasitic capacitance, e.g., a parasitic capacitance of the inductor, a drain parasitic capacitance of a MOS transistor, a gate parasitic capacitance of the MOS transistor or the like, is generated. Generation of such a parasitic capacitance is inevitable, and thus there is a problem that such a parasitic capacitance reduces a fluctuation width in a variable capacity C of the LC resonant circuit. For example, if a fluctuation amount of the capacity C is .DELTA.C, a parasitic capacitance amount is added to a denominator as a non-fluctuation portion while a design fluctuation rate is assumed to be .DELTA.C/C. Thus, in practice, there is a problem that the fluctuation rate becomes small, i.e., .DELTA.C/(C+parasitic capacitance). Though it is dependent on a circuit design or the like, if the capacity C and the parasitic capacitance are about equal, a change rate is reduced to about 1/2.

Because of the presence of such a parasitic capacitance, a ratio of the capacity value of the LC resonant circuit occupied by the variable capacity C is inevitably reduced. Thus, compared with a change rate of the variable capacity C, a change in the capacity value of the LC resonant circuit is reduced and, consequently, a variable range of an oscillation frequency is narrowed.

Today, however, a frequency band used for a portable telephone, a radio LAN device or the like has been widened, and there is a case where a plurality of frequency bands are dealt with by one device. Thus, there has been an increase in demand for expansion of a change width of the oscillation frequency. From this viewpoint, realization of a variable inductor is desired.

The variable inductor can be applied to an amplifier provided with an inductor. For example, in an amplifier which comprises the inductor for degeneration, if its inductance is reduced, distortion characteristics are deteriorated while a gain and noise characteristics of the amplifier are improved. Conversely, if the inductance is increased, a gain and noise characteristics are deteriorated while distortion characteristics are improved. Because of this trade-off relation, an inductance value is decided to obtain desired characteristics when the amplifier is designed.

Generally, realization of low distortion characteristics while maintaining a high gain and low noise characteristics is dealt with by increasing the amount of a current. In the amplifier used for a receiver of a radio terminal, characteristics necessary for the amplifier are different depending on a sized of a received signal. Generally, since it is considered important to amplify a signal at low noise if the received signal is small, a good gain and good noise characteristics are required of the amplifier. On the other hand, if the received signal is large, good distortion characteristics are required.

In the conventional amplifier which comprises the inductor for degeneration, since inductance is fixed, the amount of a supplied current is controlled in order to change the characteristics of the amplifier. That is, to improve distortion characteristics, a current is controlled so as to increase the amount of a supplied current. However, an increase in the amount of a current made to change the characteristics of the amplifier creates a problem of increased power consumption.

Further, there is disclosed a circuit example of a variable gain amplifier in Dual-Band High-linearity Variable-Gain Low-Noise Amplifier for Wireless Applications K. L. Fong, "Dual-Band High-Linearity Variable-Gain Low-Noise Amplifier for Wireless Applications," IEEE ISSCC99, pp 224 to 225, 1999. In this variable gain amplifier, a first-stage common emitter circuit constituted of a first transistor Q1 is always operated, and gain switching is realized by switching second to fourth transistors Q2 to Q4 which constitute a first-stage common base circuit. Since the first transistor Q1 is operated, input impedance is not greatly changed even if a gain is switched. However, a fixed current is always consumed, and distortion characteristics are substantially constant.

However, in the conventional circuit disclosed in Dual-Band High-Linearity Variable-Gain Low-Noise Amplifier for Wireless Applications K. L. Fong, "Dual-Band High-Linearity Variable-Gain Low-Noise Amplifier for Wireless Applications," IEEE ISSCC99, pp 224 to 225, 1999, there is a problem that a large current is consumed even when a gain is low, and distortion characteristics are about similar to those when the gain is high. To realize an amplification stage of a basically high gain and good distortion characteristics, current consumption is necessary to a certain extent. In the case of a gain which is not so high or to attenuate a signal, it is possible to realize an amplification stage of good distortion characteristics without consuming a current so greatly. However, if a plurality of amplification stages are switched, there is a problem of a change in input impedance.

As described above, the variable inductor of the conventional art has a problem in electric characteristics, and there is a problem that it is difficult to realize miniaturization, low costs and an integrated circuit.

In the radio terminal such as a portable telephone, there is a strong demand for making an adaptive characteristic change of amplifier characteristics in accordance with a received signal level, while lower power consumption is similarly demanded strongly. In the amplifier which uses a fixed inductor for degeneration, the only way to improve distortion characteristics is to increase the amount of a current, which brings about an increase in power consumption.

BRIEF SUMMARY OF THE INVENTION

An object of the present invention is to provide a variable inductor which has good electric characteristics, allows easy miniaturization and low costs, and is suited to circuit integration.

According to an aspect of the present invention, there is provided a variable inductor comprising:

a signal input terminal which receives an input signal;

a distributor including first and second active elements, configured to vary a distribution ratio of first and second currents flowing through the first and second active elements, respectively;

output terminals which output the first and second currents, respectively; and

a first inductor through which the first current flows;

a second inductor through which the second current flows and which is connected to the first inductor; and

According to an another aspect of the present invention, there is also provided an oscillator comprising:

a voltage control oscillation circuit having a first inductor; and

a frequency control circuit having a second inductor electro-magnetically coupled to the first inductor, configured to supply a control current to the second inductor, and controls an oscillation frequency of the voltage control oscillation circuit by changing the control current to change an inductance value of the first inductor.

According to an yet another aspect of the present invention, there is an amplifier comprising:

an amplifier circuit having a first inductor; and

a control section having a second inductor electro-magnetically coupled to the first inductor, configured to supply a control current to the second inductor, and change the control current to change an inductance value of the first inductor, thereby changing amplification characteristics of the amplifier circuit.

According to an further aspect of the present invention, there is an amplifier comprising:

an input terminal which receives an input signal;

a variable gain amplification circuit configured to amplify the input signal with a variable gain of an amplification, which includes a plurality of amplification stages arranged in parallel; and

an input impedance adjustment circuit including a variable resistor circuit connected to the input terminal, configured to adjust a resistance value of the variable resistor to compensate a change in input impedance in accordance with a change of the gain.

BRIEF DESCRIPTION OF THE SEVERAL VIEWS OF THE DRAWING

FIG. 1 is a circuit diagram schematically showing a variable inductor according to a first embodiment of the present invention.

FIG. 2 is a circuit diagram schematically showing a modified example of the variable inductor shown in FIG. 1.

FIG. 3 is a circuit diagram schematically showing a variable inductor according to a second embodiment of the present invention.

FIG. 4 is a circuit diagram schematically showing a modified example of the variable inductor shown in FIG. 3.

FIG. 5 is a circuit diagram schematically showing a variable inductor according to a third embodiment of the present invention.

FIG. 6 is a circuit diagram schematically showing a variable inductor according to a fourth embodiment of the present invention.

FIG. 7 is a circuit diagram schematically showing a variable inductor according to a fifth embodiment of the present invention.

FIG. 8 is a circuit diagram schematically showing a variable inductor according to a sixth embodiment of the present invention.

FIG. 9 is a circuit diagram schematically showing a variable inductor according to a seventh embodiment of the present invention.

FIG. 10 is a circuit diagram schematically showing a variable inductor according to a modification of a seventh embodiment of the present invention.

FIG. 11 is a block diagram schematically showing an oscillator according to an embodiment of the present invention.

FIG. 12 is a block diagram schematically showing an oscillator according to another embodiment of the present invention.

FIG. 13 is a block diagram schematically showing a circuit where a voltage-current conversion circuit of FIG. 12 is constituted of cascode connected transistors.

FIG. 14 is a block diagram schematically showing an oscillator according to a yet another embodiment of the present invention where a plurality of differential pairs are arranged.

FIG. 15 is a block diagram schematically showing an oscillator according to a still another embodiment of the present invention which uses a variable phase shifter and a variable gain amplifier.

FIG. 16 is a block diagram schematically showing Colpitts oscillation circuit according to a yet another embodiment of the present invention.

FIG. 17 is a plan view schematically showing a layout pattern of the inductor.

FIG. 18 is a plan view schematically showing another layout pattern of the inductor.

FIG. 19 is a plan view schematically showing yet another layout pattern of the inductor. Schematic view of the inductor explaining the embodiment of the present invention.

FIG. 20 is a block diagram schematically showing a radio terminal which comprises the oscillator shown in each of FIGS. 11 to 16.

FIG. 21 is a block diagram schematically showing an amplifier according to an embodiment of the present invention.

FIG. 22 is a circuit diagram showing a specific circuit of the differential amplifier shown in FIG. 21.

FIG. 23 is a block diagram showing an amplifier according to an another embodiment of the present invention.

FIG. 24 is a circuit diagram showing a specific circuit of the differential amplifier shown in FIG. 23.

FIG. 25 is a circuit diagram showing another specific circuit of the differential amplifier shown in FIG. 23.

FIGS. 26A and 26B are views each showing a circuit example of a variable resistor shown in FIG. 25.

FIG. 27 is an equivalent circuit diagram showing an input impedance of amplification stages shown in FIGS. 24 and 25.

FIG. 28 is a circuit diagram showing a circuit of a modified example of a circuit shown in FIG. 25.

FIG. 29 is a circuit diagram showing yet another specific circuit example of the differential amplifier shown in FIG. 21.

FIG. 30 is a circuit diagram showing yet another circuit of a modified example of the differential amplifier shown in FIG. 23.

FIG. 31 is a block diagram schematically showing a radio terminal which incorporates the amplifier described above with reference to each of FIGS. 21 to 30.

FIG. 32 is a block diagram showing basic circuitry of an amplifier which comprises an inductor having no mutual inductance according to an another embodiment of the present invention.

FIG. 33 is a block diagram showing a circuit to realize a variable resistor shown in FIG. 32.

FIG. 34 is a block diagram showing a yet another circuit to realize the variable resistor shown in FIG. 32.

FIG. 35 is a block diagram showing a circuit to realize the variable resistor shown in FIG. 34.

FIG. 36 is a graph showing a result of simulation 25 of input impedance of a portion excluding a MOSFET of an input portion in the circuit shown in FIG. 35.

FIG. 37 is a block diagram showing a yet another circuit to realize the variable resistor shown in FIG. 34.

FIG. 38 is a block diagram showing a still another circuit to realize the variable resistor shown in FIG. 34.

FIG. 39 is a block diagram showing a circuit example of a radio terminal where an amplifier circuit of FIG. 32 is applied to a low-noise amplifier of a radio equipment.

DETAILED DESCRIPTION OF THE INVENTION

Next, detailed description will be made of a variable inductor, an oscillator which incorporates the variable inductor in a circuit, a radio terminal which comprises this oscillator, an amplifier which incorporates the variable inductor in a circuit, and a radio terminal which comprises this amplifier according to the embodiments of the present invention with reference to the accompanying drawings.

The description will be made by way of example where an FET is used as a transistor which is an active element used for a distributor of a variable inductor circuit. However, a bipolar transistor can be used in place of the FET to realize the variable inductor circuit.

<Variable Inductor>

First Embodiment

First, description will be made of a variable inductor according to a basic embodiment of the present invention. FIGS. 1 and 2 show circuitry of a viable inductor of a first embodiment of the present invention. An input signal (Input) input to a signal input terminal 11 is distributed to a plurality of signal paths 13a, 13b, . . . , 13n by a distributor 12 constituted by using an active element. Inductors 14a, 14b, . . . 14b are inserted into the signal paths 13a, 13b, . . . , 13n. The inductors 14a, 14b, . . . 14n are constituted of, e.g., spirally formed leads, and arranged close to one another to be interconnected.

The inductors 14a, 14b, . . . 14n generate magnetic fluxes depending on sizes of input signals, and each magnetic flux is applied on the other close inductor to interconnect each inductor with this other inductor. Thus, the inductors 14a, 14b, . . . 14n have self inductance caused by a magnetic flux generated by each inductor itself, and mutual inductance decided by a magnetic flux generated by the other inductor coupled with each inductor. For example, inductance of the inductor 14a, i.e., inductance between terminals 15, 16 (called variable inductor terminals) of both ends of the inductor 4a is decided by self inductance Lsa of the inductor 14a, and mutual inductance Mab, . . . , Man between the inductor 14a and the other inductors 14b, . . . 14n. Here, depending on directions of the interconnected inductors (direction considering both of current and winding wire directions), the interconnected inductors can be set in a relation of reinforcing or weakening magnetic fluxes generated by the inductors. Thus, the inductance La between the variable inductor terminals 15, 16 can be set large or small with respect to the self-inductance Lsa.

The signals distributed from the distributor 12 to the signal paths 13a, 13b, . . . , 13n are supplied to the inductors 14a, 14b, . . . 14n. Here, by adjusting a distribution ratio of the signals distributed from the distributor 12 to the inductors 14a, 14b, . . . 14n, the mutual inductance Mab, . . . , Man, i.e., the amount of a magnetic flux by interconnection of the inductors 14a, 14b, . . . 14n, can be controlled. Thus, the inductance between the variable inductor terminals 15, 16 can be set to a desired value.

The distributor 12 is constituted of an active element such as a transistor as described later. Accordingly, different from the conventional variable inductor circuit which uses the directional coupler, the distributor 12 shown in FIGS. 1 and 2 can be easily formed into an integrated circuit. In the circuit shown in FIG. 1, the distributor 12 is connected between the signal input terminal 11 and one end of each of the inductors 14a, 14b, . . . 14n. In the circuit shown in FIG. 2, one end and the other end of each of the inductors 14a, 14b, . . . 14n are connected to the distributor 12.

Next, some more specific embodiments of the variable inductor shown in FIGS. 1 and 2 will be described.

Second Embodiment

FIGS. 3 and 4 show a variable inductor where the distributor 12 includes a common source circuit according to the embodiment. The circuit of this embodiment is equivalent to the specific circuit example of the basic circuitry shown in FIG. 1. The input signal from the signal input terminal 11 is amplified by a common source circuit, e.g., MOSFET's (simply referred to as transistor, hereinafter) 21a, 21b to be distributed to two signal paths, and supplied to the inductors 14a, 14b inserted into these signal paths. Gate terminals of the transistors 21a, 21b are connected to the signal input terminal 11, source terminals are connected to ends of the inductors 14a, 14b, and drain terminals are connected to output terminals 22a, 22b of the variable inductor. In this variable inductor, signal currents amplified by the MOSFET's 21a, 21b are outputted from output terminals 22a, 22b of the variable inductor.

The other ends of the inductors 14a, 14b are connected to ends of power sources 23a, 23b and ends of capacitors 24a, 24b, and grounded with respect to AC components (high frequency components). The other ends of the power sources 23a, 23b and the other ends of the capacitors 24a, 24b are connected to the ground. Among currents flowing through the inductors 14a, 14b, DC components flow through the power sources 23a, 23b, and AC components (high frequency components) are bypassed by the capacitors 24a, 24b.

In the variable inductor shown in FIGS. 3 and 4, black points given near symbols indicating the inductors 14a, 14b represent winding start positions of the inductors. In the inductors where these positions are the same, directions of all the winding wires thereof are the same, and phases of magnetic fluxes are the same.

If the directions of the inductors 14a, 14b are the same as shown in FIG. 3, magnetic flues are generated in reinforcing directions in the inductors 14a, 14b. Thus, if self inductance of the single inductor 14a is Lsa, self inductance of the single inductor 14b is Lsb, and an coupling coefficient between the inductors 14a, 14b is kab, effective inductance of the inductor 14a considering interconnection, i.e., inductance La between the variable inductor terminals 15, 16 is represented by the following equation (1): La=lsa+kabLsb (1)

If mutual inductance between the inductors 14a, 14b is Mab, the equation (1) is represented by the following equation (2): La=Lsa+Mab (2)

On the other hand, if the directions of the inductors 14a, 14b are opposite as shown in FIG. 4, magnetic fluxes are generated in a weakening relation in the inductors 14a, 14b. This inductance La is represented by the following equation (3) or (4): La=Lsa-kabLsb (3) La=Lsa-Mab (4)

Here, the coupling coefficient kab, i.e., mutual inductance Mab, is decided based on a physical arrangement of the inductors 14a, 14b, sizes of the transistors 21a, 21b, current values Ia, Ib or the power sources 23a, 23b etc. Thus, even without changing the physical arrangement of the inductors 14a, 14b, by adjusting the current values Ia, Ib or the sizes of the transistors 21a, 21b, a value of the inductance La can be varied.

For example, the current source 23b is set as a variable current source which can control the current value Ib by a control signal from the outside, and this current value Ib is continuously changed. Accordingly, the mutual inductance Mab is changed. Thus, the inductance La between the variable inductor terminals 15, 16 is continuously changed. When the power source 23b is turned ON/OFF, the inductance La can be switched between (Lsa+Mab) or (Lsa-Mab) and Lsa in a binary manner.

According to the embodiment shown in FIGS. 3 and 4, the common source circuit by the FET is used for the distributor 12. However, a common emitter circuit by a bipolar transistor can also be used. In the case of the emitter-grounded circuit, the gate terminal, the drain terminal and the source terminal of the FED are replaced by a base terminal, a collector terminal and an emitter terminal of the bipolar transistor. In the circuit shown in FIGS. 1 and 2, the two inductors are used. However, even for a circuit which uses three or more inductors, a constitution similar to the embodiment can be applied.

Moreover, as a modified example, the distributor 12 may be constituted of a plurality of transistors where gate or base terminals of distributors are connected in common to the signal input terminal 11 through at least one inductor of the plurality of inductors. As another modified example of the embodiment, the distributor 12 may be constituted of a plurality of transistors where gate or base terminals are connected in common to the signal input terminal 11, and drain or collector terminals are connected to ends of the plurality of inductors.

Third Embodiment

FIG. 5 shows a variable inductor according to a third embodiment which uses a transistor circuit where a gate is grounded to a distributor 12. A source terminal of at least one of first transistors 31a, 31b, . . . , 31n, and source terminals of a plurality of second transistors 32a, 32b, . . . , 32n are connected to a signal input terminal 11. Drain terminals of the first transistors 31a, 31b, . . . , 31n are connected to one end of a first inductor 14a, and gate terminals are connected to control signal input terminals 33a, 33b, . . . , 33n. Drain terminals of the second transistors 32a, 32b, . . . , 32c are connected in common to one end of a second inductor 14b, and gate terminals are connected to control signal input terminals 34a, 34b, . . . , 34n.

Control signals .phi.33a, .phi.33b, . . . , .phi.33n are input to the control signal input terminals 33a, 33b, . . . , 33c. Control signals .phi.34a, .phi.34b, . . . .phi.34n are input to the control signal input terminals 34a, 34b, . . . , 34c. When the control signals .phi.33a, .phi.33b, . . . , .phi.33n and .phi.34a, .phi.34b, . . . , .phi.34b are changed in a binary manner, a distribution ratio of signal levels to the inductors 14a, 14b, i.e., a ratio of currents flowing through the inductors 14a, 14b, is changed. Thus, mutual inductance Mab between the inductors 14a, 14b is changed and, as a result, effective inductance La (inductance between terminals 15, 16) of the inductor 14a can be changed.

A distribution ratio of a signal level to the inductor 14a is decided based on the number of transistors turned ON by the control signals .phi.33a, .phi.33b, . . . , .phi.33n among the transistors 31a, 31b, . . . , 31c. Similarly, a distribution ratio of a signal level to the inductor 14b is decided based on the number of transistors turned ON by the control signals .phi.34a, .phi.34b, . . . , .phi.34n among the transistors 32a, 32b, . . . , 32c.

The inductance La between the terminals 15, 16 may be continuously changed by setting the control signals .phi.33a, .phi.33b, . . . , .phi.33n, and .phi.34a, .phi.34b, . . . , .phi.34n as analog signals and continuously changing currents flowing through the transistors 31a, 31b, . . . , 31n and the transistors 32a, 32b, . . . , 32c.

In FIG. 5, the plurality of transistors 31a, 31b, . . . , 31n are connected to the inductor 14a. However, if a small variable range of inductance is allowed, the number of first transistors may be one. Similarly, the plurality of second transistors 32a, 32b, . . . , 32n are connected to the inductor 14b. However, if a small variable range of inductance is allowed, the number of second inductors may be one.

In the variable inductor of the embodiment, the common gate circuit by the FET is used for the distributor 12. However, a common base circuit by a bipolar transistor can also be used. In the case of the base-grounded circuit, the gate terminal, the drain terminal and the source terminal of the FED are replaced by a base terminal, a collector terminal and an emitter terminal of the bipolar transistor.

Moreover, as a modified example, the distributor 12 may be constituted of at least one first transistor where a source or emitter terminal is connected to the signal input terminal 11 through at least one first inductor, and a gate or base terminal is connected to the control signal input terminal, and at least one second transistor where a source or emitter terminal is connected to the signal input terminal 11 through the first inductor, a drain or collector terminal is connected to one end of at least one second inductor coupled to the first inductor, and a gate or base terminal is connected to the control signal input terminal.

Fourth Embodiment

FIG. 6 shows a variable inductor according to a fourth embodiment of the present invention which uses a source follower circuit for a distributor 12. Gate terminals of a plurality of transistors 41a, 41b (two in the example of FIG. 6) are connected to a signal input terminal 11. Drain terminals of the transistors 41a, 41b are connected to a power source Vdd which is a constant potential point, and source terminals are connected to ends of inductors 14a, 14b. Further, current sources 43a, 43b are connected to the source terminals of the transistors 41a, 41b. Thus, the transistors 41a, 41b operate as source follower circuits.

Here, as in the case of the second embodiment, if the current source 43a is set as a variable current source which can control a current value Ib by a control signal from the outside, and this current value Ib is continuously changed, mutual inductance Mab between the inductors 14a, 14b is accordingly changed, whereby inductance La between the terminals 15, 16 is continuously changed. When the current source 43b is turned ON/OFF, the inductance La is switched between (Lsa+Mab) or (Lsa-Mab) and Lsa in a binary manner where self-inductance of the inductor 14a is set as Lsa.

According to the embodiment, the source follower circuit by the FET is used for the distributor 12. However, it is apparent that an emitter follower circuit by a bipolar transistor can also be used.

Further, as a modified example of the embodiment, the distributor 12 may be constituted of a plurality of transistors where gate or base terminals are connected to the signal input terminals 11 through the inductors, and the drain or collector terminals are connected to the constant potential point.

Fifth Embodiment

FIG. 7 shows a variable inductor of a fourth embodiment of the present invention which uses cascode-connected circuits for a distributor 12. Gate terminals of first and second transistors 51, 52 are connected to a signal input terminal 11. Source terminals of the transistors 51, 52 are connected to current sources 53, 54. A drain terminal of the first transistor 51 is connected to a source terminal of a third transistor 55, and a drain terminal of the second transistor 52 is connected in common to source terminals of a plurality of fourth transistors 56a, 56b (two in the example of FIG. 7). That is, the transistor 51 and the transistor 55 are cascode-connected, and the transistor 52 and the transistors 56a, 56b are cascode-connected.

Drain terminals of the third transistor 55 and the fourth transistors 56a, 56b are connected to ends of inductors 14a, 14b, 14c. A gate terminal of the third transistor 55 is connected to a control signal input terminal 58, and gate terminals of the fourth transistors 56a, 56b are connected to control signal input terminals 57a, 57b. As indicated by black points given near symbols of the inductors 14a, 14b, 14c, the inductor 14b is arranged in a magnetic flux reinforcing direction, and the inductor 14c is arranged in a magnetic flux weakening direction with respect to the inductor 14a.

Now, in a state where a control signal .phi. is input to the control signal input terminal 58 to turn ON the transistor 55, if a control signal .phi.+ is input to the control signal input terminal 57a and a control signal .phi.- is input to the control signal input terminal 57b to turn ON the transistor 56a connected to the inductor 14b and turn OFF the transistor 56b connected to the inductor 14c, inductance La between both ends 15, 16 of the inductor 14a becomes larger than self inductance Lsa of the inductor 14a. Conversely, if a control signal .phi.- is input to the control signal input terminal 57a and a control signal .phi.+ is input to the control signal input terminal 57b to turn OFF the transistor 56a and turn ON the transistor 56b, inductance La between both ends 15, 16 of the inductor 14a becomes smaller compared with self inductance Lsa.

Thus, by using the cascode-connected circuits for the distributor 12, the fourth transistors 56a, 56b cascode-connected to the second transistor 52 are turned ON/OFF to selectively supply signals to the inductors 14b, 14c. As a result, the inductance La between the terminals 15, 16 can be increased/decreased.

According to the embodiment, the cascode-connected circuits by the FET are used for the distributor 12. However, it is apparent that cascode-connected circuits by a bipolar transistor can also be used.

Sixth Embodiment

FIG. 8 shows a variable inductor according to a fifth embodiment of the present invention. A gate terminal of a transistor 61 which constitutes an amplifier 60 is connected to a signal input terminal 11. A current source 63 and an AC (high frequency) bypass capacitor 64 are connected to a source terminal of the transistor 61. One end of an inductor 14a and an input terminal of a buffer circuit 62 are connected to a drain terminal of the transistor 61, and one end of the other inductor 14b is connected to an output terminal of the buffer circuit 62.

The buffer circuit 62 is constituted of a circuit formed on an integrated circuit, e.g., a common source circuit or a source follower circuit, and its gain is varied. By changing a gain of the buffer circuit 62, a distribution ratio of signal levels to the inductors 14a, 14b is changed as in the cases of the aforementioned embodiments. Accordingly, mutual inductance is changed to enable a change in inductance between variable inductor terminals 15, 16.

According to the embodiment, the FET is used for the transistor 61 in the distributor 12. However, as in the cases of the aforementioned embodiments, a bipolar transistor may be used.

Seventh Embodiment

The embodiments have been described by way of case where the distributors 12 are all in single-end circuitry. However, the distributor 12 may be constituted of a differential circuit. As a variable inductor of a seventh embodiment of the present invention, an example where the distributor 12 is constituted of a differential circuit is shown in FIG. 9. In the circuit shown in FIG. 9, a differential amplifier cascode-connected to the distributor 12 is used.

Gate terminals of differential pair transistors 71a, 71b and gate terminals of differential pair transistors 72a, 72b are connected to differential input terminals 11a, 11b. A source terminal of the differential pair transistors 71a, 71b is connected to a current source 78, and drain terminals of the differential pair transistors 72a, 72b are connected to a current source 79. Drain terminals of the differential pair transistors 71a, 71b are connected to source terminals of transistors 73a, 73b, and the drain terminals of the differential pair transistors 72a, 72b are connected to source terminals of transistors 74a, 74b and source terminals of transistors 75a, 75b.

Gate terminals of the transistors 73a, 73b, 74a, 74b, 75a, 75b are connected to a control signal input terminal. Drain terminals of the transistors 73a, 73b are connected to differential output terminals 22a, 22b, and to ends of inductors 76a, 76b. Drain terminals of the transistor 74a and the transistor 75b are connected in common to one end of an inductor 77a. Drain terminals of the transistor 74b and the transistor 75a are connected in common to one end of an inductor 77b. The other ends of the inductors 76a, 76b, 77a, 77b are connected to a power source Vdd which is a constant potential point.

Signals Input+, Input- opposite to each other in phase are input to the differential input terminals 11a, 11b. The entry of the signals Input+, Input- opposite to each other in phase is equivalent to reversal of codes of mutual inductance M12 between the inductors 76a, 77a and mutual inductance M13 between the inductors 76b, 77b. Accordingly, by control signals .phi.+ and .phi.-, interconnection for mutual inductance M12 and M13 can be controlled in magnetic flux reinforcing direction and in a magnetic flux weakening direction. Thus, the inductor can be changed.

FIG. 10 shows a distributor provided with a differential circuit according to a modification of the circuit configuration shown in FIG. 9, in which bipolar transistors are used instead of the FETs. The distributor shown in FIG. 10 can be operated in a same manner as that of FIG. 9, even if the different type transistors are incorporated in the distributor.

As described above, according to the variable inductor of the embodiment of the present invention, it is possible to electro-magnetically change the inductor by using the distributor to be formed on the integrated circuit. Moreover, such a variable inductor is suited for integrated circuit formation because of good electric characteristics, easy miniaturization and easy achievement of low costs.

<Oscillator Comprising Variable Inductor and Radio Terminal Comprising This Oscillator>

Next, description will be made of an oscillator which incorporates the aforementioned variable inductor of the present invention, and a radio terminal which comprises this oscillator.

FIG. 11 is a block diagram schematically showing an oscillator according to an embodiment of the present invention.

The oscillator shown in FIG. 11 comprises a core circuit section (VCO core) 102 which includes an LC resonant circuit having an inductor L0, and an oscillation frequency control section 104-1 (frequency controller) which has an inductor L11 to be electro-magnetically coupled to the inductor L0 based on a coupling coefficient k1 and can control a current supplied to the inductor L11.

A control signal Control_1 is input to the oscillation frequency control section 104-1 and, in accordance with this control signal Control_1, at least one or both of an amplitude and a phase of a current flowing through the inductor L11 is changed. As a result, inductance of the inductor L0 of the core circuit section 102 coupled to the inductor L11 is changed to cause a change in an oscillation frequency. For example, when a current of a direction where a magnetic flux is generated in a reinforcing direction with a magnetic flux generated in the inductor L0 flows through the inductor L11, a value of inductance of the inductor L0 becomes large, and an oscillation frequency becomes small in proportion to -1/2 square of the value of the inductance. When a current of a direction where a magnetic flux generated in the inductor L0 is weakened flows through the inductor L11, a value of inductance of the inductor L0 becomes small and, as a result, an oscillation frequency becomes large. By setting the currents flowing through the inductor L0 and the inductor L11 in the same direction to control current amplitudes, an increase/decrease of the inductor L0 can be controlled.

Generally, a control range of a frequency obtained by varying a capacity C is about C max/C min=2. On the other hand, in the system of changing inductance, for example, if currents flowing through the inductor L and the inductor L11 are set to equal values, and k=about 0.7 is set, L min=(1-k)L, L max=(1+k) are established to realize a large fluctuation range of L max/L min=about 6.

Normally, to change an oscillation frequency by a variable capacity, only a change in a range of 5 to 10% of an oscillation frequency is obtained. However, in the oscillator, which comprises the variable inductor as shown in FIG. 11, an oscillation frequency can be changed to a range of 50% to 100%.

The number of inductors L11 is not limited to one. As shown in FIG. 11, a plurality of inductors L11 to L1n, e.g., inductors L11, may be disposed. In the circuit shown in FIG. 11, n pieces of oscillation frequency control sections 104-n are disposed, and inductors L11 to L1n of the oscillation frequency control sections are electro-magnetically coupled to the inductor L of the VCO circuit 102 based on coupling coefficients (k1 to kn).

If it is only one inductor L11 that is electro-magnetically coupled to one oscillation frequency control section 104-1, i.e., the inductor L of the core circuit section 102, by changing an amplitude or a phase of a current flowing through the inductor L11, it is possible to change an inductance value of the inductor L.

In the circuit which comprises the plurality of inductors L11 to L1n, the VCO circuit 102 can be controlled on various modes. For example, even if a coupling coefficient is equal to that of the inductor L11 and a current identical to that which flows through the inductor L11 flows through the inductor L1n, the number of inductors L1n through which the current is turned ON/OFF to flow may be changed to change the activated inductor L1n electro-magnetically coupled to the inductor L.

The inductors L11 to L1n having different coupling coefficients k1 to kn can be prepared to be switched to be used. Further, currents flowing through the individual inductors L11 to L1n are changed to enable more meticulous control of the core circuit section 102.

FIG. 12 shows a circuit example where an oscillation signal (voltage signal) is input from a voltage control oscillation circuit 106 of a differential LC resonant type equivalent to the VCO core circuit 102 to the oscillation frequency control circuit 104, and this oscillation signal is subjected to voltage-current conversion to be supplied to the inductors L1, L2 coupled to the inductors L01, L02 of the core circuit section 102.

In the circuit shown in FIG. 12, one end of the inductor L01 is connected to the current source having a power supply voltage Vdd, and the other end is connected to a variable capacitor VC1 which comprises a drain and a diode of a MOS transistor T1. A source of this MOS transistor T1 is grounded.

Similarly, one end of the inductor L02 is connected to the current source having the power supply voltage Vdd, and the other end is connected to a variable capacitor VC2 which comprises a drain and a diode of a MOS transistor T2. A source of this MOS transistor T2 is grounded.

The drain of the MOS transistor T1 is connected to a gate of the MOS transistor T2, and similarly a drain of the MOS transistor T2 is connected to a gate of the MOS transistor T1.

A capacity control voltage Vctr1 is supplied to the variable capacitors VC1, VC2 to decide capacities thereof. A resonance frequency is determined in accordance with parallel connections (L1-VC1), (L20VC2) of the variable capacitors VC1, VC2 and the inductors L1, L2.

In FIG. 12, the variable capacitors VC1, VC2 are used. However, fixed capacity capacitors may be used. In the case of using the fixed capacity capacitors, only a fluctuation portion of frequency control dependent on the variable capacitors VC1, VC2 is eliminated, and there are no changes in operation.

From the oscillator 106, an output 1 (Output_1) from the drain of the MOS transistor T1 and an output 2 (Output_2) from the drain of the MOS transistor T2 are outputted.

In order to change an oscillation frequency of the output signal, the output 1 and the output 2 (Output_1, Output_2) are input to a current-voltage conversion circuit (V-I converter) 108 of the oscillation frequency control circuit 104 to control currents flowing through the inductor L1 electro-magnetically coupled to the inductor L01 based on a coupling coefficient k1 and through the inductor L2 electron-magnetically coupled to the inductor L02 based on a coupling coefficient k2.

Here, it is assumed that k1=0.7, k2=0.7 are set, inductance values of the inductors (L01, L02, L1, L2) are equal at L, and currents of equal sizes flow through the inductors.

If a current between the electro-magnetically coupled inductors is in a magnetic field reinforcing direction, an inductance value of the inductor L01 is increased from L0 to 1.7L0.

If a current flows in a magnetic field weakening direction, an inductance value of the inductor L01 is reduced to 0.3L0.

Accordingly, it can be understood that if a lower part of an oscillation frequency is near fLo=2 GHz, a higher part thereof becomes fHi=(0.3/1.7)-1/2fLo, and thus an oscillation frequency of 4 GHz or higher can be obtained.

FIG. 13 shows a specific example where the voltage-current conversion circuit 108 is constituted of cascode-connected transistors. A voltage control oscillation circuit 16 of a differential LC resonant type is similar to the circuitry shown in FIG. 12, and denoted by the same reference numeral in FIG. 13, and thus description thereof will be omitted. Hereinafter, a frequency control section 104 will be described.

A drain of a transistor T12 is connected to the inductor L1, and a source of the transistor T12 is connected to a drain of a transistor T11 to which gate an output (Output_1) is input. A source of the transistor T12 is grounded through a current source 110. A source of a transistor T13 which drain is connected to the inductor L2 is connected to the drain of the transistor T11. A control signal .phi.+ is input to a gate of the transistor T12, and a control signal .phi.- is input to a gate of the transistor T13. That is, the inductor L1 is connected to the transistor T12 cascode-connected to the transistor T11 where the source is grounded through the current source and the output 1 (Output_1) is supplied to the gate, and the control signal .phi.+ is supplied to the gate of the transistor T12. The inductor L2 is connected to the transistor T13 cascode-connected to the transistor T11, and the control signal .phi.- is supplied to the gate of the transistor T13.

Similarly, a drain of a transistor T15 is connected to the inductor L2, and a source of the transistor T15 is connected to a drain of a transistor T14 to which gate an output (Output_2) is input. A source of the transistor T14 is grounded through a current source. A source of a transistor T16 which drain is connected to the inductor L1 is connected to the drain of the transistor T14. A control signal .phi.+ is input to a gate of the transistor T15, and a control signal .phi.- is input to the transistor T16. That is, the inductor L2 is connected to the transistor T15 cascode-connected to the transistor T14 where the source is grounded through the current source and the output 2 (Output_2) is supplied to the gate, and the control signal .phi.+ is supplied to the gate of the transistor T15. The inductor L1 is connected to the transistor T16 cascode-connected to the transistor T14, and the control signal .phi.- is supplied to the gate of the transistor T16.

By changing the control signals .phi.+, .phi.-, for example, directions of currents flowing through the inductors L1, L2 can be reversed. Additionally, by properly setting potentials of the control signals .phi.+, .phi.-, amplitudes of the currents flowing through the inductors L1, L2 can be changed.

Thus, by controlling the control signals .phi.+, .phi.-, the currents flowing through the inductors L1, L2 are controlled and, as a result, it is possible to control inductance values of the inductors L01, L02 coupled to the inductors L1, L2. For example, when a current flowing through the inductor L01 (L02) and a current flowing through the inductor L1 (L2) are in-phase, an inductance value of the inductor L01 (L02) becomes large and, as a result, an oscillation frequency becomes low. When a current flowing through the inductor L01 (L02) and a current flowing through the inductor L1 (L2) are reverse in phase, an inductance value of the inductor L01 (L02) becomes small and, as a result, an oscillation frequency becomes high.

FIG. 14 shows an oscillator of an embodiment where a plurality of differential pairs are arranged.

One end of an inductor L01 is connected to a current source having a power supply voltage Vdd, and the other end is connected to a drain of a MOS transistor T1 and a capacitor C1. A source of this MOS transistor T1 is grounded through a current source I0.

Similarly, one end of an inductor L02 is connected to a current source having a power supply voltage Vdd, and the other end is connected to a drain of a MOS transistor T2 and a capacitor C2 constituted of a drain. A source of this transistor T2 is grounded through the current source I0.

The drain of the transistor T1 is connected to a gate of the transistor T2 and, similarly, the drain of the transistor T2 is connected to a gate of the transistor T1.

In the circuit shown in FIG. 14, capacitances of the capacitors C1, C2 are fixed. However, variable capacitors may be used as in the case shown in FIG. 12.

In the circuit shown in FIG. 14, an oscillation frequency control section is constituted of a plurality of differential pairs. An inductor L1n one end of which is connected to the power supply voltage Vdd, e.g., an inductor L11, is electro-magnetically connected to an inductor L01 of a VCO circuit 106 based on a coupling coefficient k1n, e.g., a coupling coefficient k11, and the other end is connected to a MOS transistor T1n, e.g., a drain of a transistor T11. A capacitor C1(n), e.g., a capacitor C11, is branched to be connected between a drain of the transistor T1n and the inductor L1n. An output 1 (Output_1) of the VCO circuit is supplied to a gate of the transistor T1n, and a source of the transistor T1n is grounded through a variable current source In, e.g., a variable current source I1.

Similarly, an inductor L2n one end of which is connected to the power supply voltage Vdd, e.g., an inductor L21, is electro-magnetically coupled to an inductor L02 of the VCO circuit based on a coupling coefficient k2n, e.g., a coupling coefficient k21, and the other end is connected to a MOS transistor T2n, e.g., a drain of a transistor T21. A capacitor C2(n), e.g., a capacitor C21, is branched to be connected between a drain of the transistor T2n and the inductor L2n.

An output 2 (Output_2) of the VCO circuit 106 is supplied to a gate of the transistor T2n, and a source of the transistor T2n is grounded through the variable current source In.

For the capacitor C1n and the capacitor C2n, e.g., the capacitor C11 and the capacitor C21, the other ends connected to the inductors are interconnected.

A plurality of such differential pairs are prepared, and current values of variable current sources I0 to In are changed or turned ON/OFF to change inductance of the inductors L0, L02 of the VCO circuit 106. Accordingly, it is possible to control an oscillation frequency.

FIG. 15 shows an oscillator of an embodiment which uses a variable phase shifter and a variable gain amplifier.

A VCO circuit 106 has components similar to those shown in FIG. 12, denoted by similar reference numerals, and description thereof will be omitted. Buffer circuits 112, 113 having large impedance seen from the outside are connected to an output terminal.

Output signals (Output_1, Output_2) from the oscillator are input into variable phase shifters 116, 118 of an oscillation frequency control circuit 104. The phases of the input signals are properly controlled by variable phase shifters 120, 122 and are output from the shifters 120, 122. Subsequently, a





DISCLAIMER

Snappy Quick is a free service that searches the Internet and indexes all files found so that you may search quickly and easily for shared files. These files are created and made available individually by users whose identity we are not aware of and who we have no control over. In essence we function like a search engine tool; these files ARE NOT STORED OR SERVED BY OUR NETWORK. We are not responsible for any materials obtained by using our service. We do not monitor any of the contents of these files. These files may contain viruses, illegal materials, materials inappropriate for minors, offensive files and the like. BY USING OUR SERVICE, YOU ASSUME FULL RESPONSIBILITY FOR DOWNLOADING THESE MATERIALS AND WILL INDEMNIFY US FOR ANY DAMAGES THAT MAY BE INCURRED.

For More Specific Information VIEW OUR TERMS OF SERVICE.

Thank you and Enjoy!